Wireless Resonance Coupled Energy Transmission

ABSTRACT

In a first aspect of the present invention, a wireless power transmission link is proposed, which while substantially maintaining resonant coupling condition (resonance frequency of the source resonant circuit is substantially equal to the resonance frequency of the load resonant circuit) detects a coupling condition of the wireless power transmission link. 
     In a further aspect of the present invention, a wireless power transmission link is suggested, which while substantially maintaining resonant coupling condition (resonance frequency of the source resonant circuit is substantially equal to the resonance frequency of the load resonance circuit) controlling the operating state of the wireless power transmission link such, that the coupling condition of wireless power transmission link is substantially limited to the critical coupling condition.

FIELD OF THE INVENTION

The current invention relates to wireless energy transmission by meansof inductive or capacitive coupled resonant circuits.

Further, the current invention relates to the control of the resonancefrequency, to control the power to be transmitted, and to electricallycontrol the coupling of coupled resonant circuits.

BACKGROUND OF THE INVENTION

Wireless power supply devices can be realized by means of inductiveand/or capacitive proximity coupling. This is used in many RFID systemsand wireless battery chargers. In this case, a source unit generates analternating electromagnetic field. This alternating electromagneticfield is coupled through coupled coils (inductive coupling) or by anopen capacitor (capacitive coupling) to a load, in the followingreferred to as a load unit. With increasing distance from the sourceunit to the load unit decreases the coupling strength and reduces thereceivable amount of power at the load unit. In the case of an opencapacitor thereby minimizing the coupling capacity, and in the case ofcoupled coils thereby increasing the leakage inductance. It is knownthat this effect can be compensated in case of the leakage inductance bya compensation capacitor and in the case of the coupling capacitor by acompensation inductor. This results in at least one resonant circuit atthe source unit side and at least one resonant circuit at the load unitside of the power transmission system. These resonant circuitscompensate the leakage inductance and coupling capacitance under thecondition that the resonant circuits are tuned exactly to the sameresonance frequency and the source unit of the power transmission systemoperates at this resonance frequency. Such coupled resonant circuits arethe base of band filters and have been used for many years for e.g. thecoupling of amplifier stages etc.

In “Wireless Power Transfer System Description” of the “Wireless PowerConsortium (WPC)”, a resonant circuit is shown, which is driven by agenerator. Here, a plurality of part inductors is selectively used in aresonant circuit in order to concentrate the radiated energy field tothe surface where load units are placed. Further, the power supply iscontrolled by an upstream voltage or current regulator.

The general disadvantage of wireless power principles, based on coupledresonant circuits in the source- and/or load unit, is the resonancefrequency detuning due to component tolerances, component aging,coupling and load changes. This detuning effect is undesirable becausethe impedance of the resonant circuit is frequency selective and apredetermined operation frequency does no longer coincide with theresonant frequency of the circuit. Consequently, the power transmissionsystem operates no longer toward a real load resistance, but also towardan inductive or capacitive component. Thus, the resulting reactive powerincreases the power dissipation. This lowers the overall efficiency inthe source unit (resonant circuit driver stage, etc.) and thus reducesthe efficiency of the entire power transmission link. In addition,distortions increase, because the driver circuits generate moreharmonics.

The known network sensing method measures the resonance frequency of thenetwork during a time interval and operates the system at this resonancefrequency thereafter, but the system has no ability to control theresonance frequency of the resonant circuit or network actively. Thiswould be very desirable, due to guidelines such as EN300330, REC7003 andITU-RSM2123, which determines maximum power levels over frequency ranges(e.g. 119 . . . 135 kHz).

Furthermore, there exist national specific constraints for narrowfrequency ranges within a frequency band that require much lower powerlevel limits. It is therefore important to control not only the powerlevel but also the spectral position of the emitted power.

Another problem relates to the product of the variable coupling (k) andthe quality (Q) in the further description referred to as energycoupling (k·Q). The quality factor Q is a measure of the energy storedin the system and the energy transferred by the system, or in otherwords a measure of the reactive power circulating in the resonantcircuit and output power of the resonant circuit. The coupling (k) isessentially determined by the geometry of the wireless coupling linksuch as distance (area, distance), its angular orientation and thecoupling medium. A change in the load resistor in a resonant circuitwill also change the energy coupling (k·Q). The system is more or lessdamped and the energy coupling (k·Q) is consequently smaller or larger.A desired constant output voltage or a desired constant output currentof the load unit requires therefore the control of the output power inthe source unit by means of a data-load modulation link and/or thecontrol of the output voltage and/or current on the load unit side.

“A Frequency Control Method for Regulating Wireless Power to ImplantableDevices” proposes frequency detuning in the source unit for the outputvoltage regulation. The problem with this solution is the detuning ofthe resonance frequency that also simultaneously controls the power ofother coupled load units if more than one load unit is used in thewireless power transmission link. An independent control is notpossible. Thanks to the performed detuning on the load side, severalload units are independently adjustable. However, disadvantageously, thesource unit is no longer loaded with a real load resistance in thewireless power transmission link in both variants. The energy transferis no longer based on coupled resonant circuits and the overall networkdoes not operate in the real domain (resonance case). Consequently, thiscauses higher losses in the source unit and/or in the load unit.

In “Wireless Power Supply for Implantable Biomedical Device Based onPrimary Input Voltage Regulation”, the output voltage of the load unitis digitized and transmitted to the source unit. The operating voltageof the source unit is controlled based on the received data of the loadunit. This approach is pursued in the standard of “Wireless PowerConsortium (WPC)”, wherein a wireless power transmission link up to 5watts is specified. This approach is efficient because only the amountof power is transmitted as needed. Unfortunately resonance detuning isnot considered.

Another fundamental problem is not considered in all known systems, andrelates to the upper energy coupling boundary value (k·Q=1), wherein themaximum possible power can be transmitted without any substantialfrequency detuning or without substantially bandwidth increase. Thiscase is very important because a small coupling factor (k) can becompensated with a higher quality (Q). In this manner, the energycoupling (k·Q) can be held constant. Thus, for example, the powertransmission distance can be increased.

In practice, however, often results in a dynamic coupling (time-varyingcoupling (k)) and/or dynamic load. Examples are different wireless powertransmission links with dynamic coupling due to variable geometries,and/or variable distance between the source unit and load unit, and/ormodifying the coupling media and especially by altering the loadresistance.

It is an essential desire to design a wireless power transmission systemas universal as possible and ideally approximating a wired connection asmuch as possible. This would automatically result in the best efficiencyand additionally serves maximal flexibility.

It is further desirable that for a given source unit the operationaldistance of the power transmission link can be optimized in designingthe load unit without any design modifications in the source unit (coilsand/or capacitors). E.g., a load unit featured with a small coupling (k)(small coupling surface, etc.) can compensate its limited range with agreater quality factor (Q). This would meet requirements of an openflexible standard, wherein only the basics shall be specified.

Additionally, multiple arbitrary load units coupled under changingcoupling conditions (changing coupling factors (k) and/or changingqualities (Q)) shall maintain operable in parallel on a source unit.

Further, it should be possible to specify the transmission frequency, asthere are radiation limits by law (amplitude and frequency), such asEN300330, REC7003 and ITU-RSM2123.

Further it should be possible to vary the transmission frequency inorder to reduce the spectral peak power.

The following invention describes methods and their implementationdetails that meet the mentioned requirements. The methods described inthe following invention and its detail implementations are featured bylow-cost implementations, stable operation and high efficiency.

BRIEF DESCRIPTION OF DRAWINGS

FIGS. 1a . . . 1 g show basic simplified equivalent circuits of a nearfield wireless power transmission system using coupled resonant circuitsand their transmission properties.

FIG. 2 shows the block diagram of a wireless power transmission systembased on resonance coupling according to a first embodiment of thecurrent invention.

FIG. 3 shows the block diagram of an wireless power transmission systembased on resonance coupling according to a second embodiment of thecurrent invention.

FIG. 4 shows waveforms of output signals of over coupling detectors inaccordance with FIGS. 2 and 3.

FIGS. 5a . . . 5 h show universal embodiments of one coupling inductanceused in two main applications according to the current invention.

FIGS. 6A and 6B show waveforms and a block diagram of a communicationcontroller according to FIGS. 2 and 3.

FIG. 7 illustrates a package structure or communication protocolaccording to FIGS. 2 and 3.

FIG. 8 shows a detailed circuit according to the current invention.

FIGS. 9A and 9B show waveforms according to the FIG. 8.

FIG. 10 shows a further detailed circuit according to the currentinvention.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1a shows a wireless power transmission system. A source couplesenergy via two coupled coils L1 and L2 to a load RL. A leakagecompensation network is implemented on the source- and load unit side tocompensate the leakage inductance.

FIG. 1b shows the wireless power transmission link according to FIG. 1awith a simplified transformer equivalent circuit 11 and the capacitorsC1 and C2 as leakage compensation. C1 and C2 form together with theprimary and secondary inductances L1 and L2 the coupled resonantcircuits having substantially the same resonance frequencies. C1 istuned with L1 to a primary resonance frequency. C2 is tuned with L2 to asecondary resonance frequency. A parallel resonant circuit, formed by C1and L1 on the primary side and a series resonant circuit formed by C2and L2 on the secondary side couple a current source I1 to a loadresistor RL. The primary leakage inductance 12 with a value of L1−M, thesecondary leakage inductance 13 with the value of L2−M and the maininductance 14 with the value M correspond to the coupled coils. Thevalues M, coupling k, primary inductance L1 and secondary inductance L2are related by the equation M=k·√{square root over (L1·L2)}. Thesecondary quality (Qs) is determined by

${{Qs} = \frac{{\omega \cdot L}\; 2}{RL}},$

wherein ω is the angular frequency. As smaller RL is, the greater is Qs.

FIG. 1c shows the wireless transmission link according to FIG. 1a with asimplified transformer equivalent circuit 11 and the capacitors C1 andC2 as leakage compensation. C1 and C2 form together with the primary andsecondary inductances L1 and L2 the coupled resonant circuits havingsubstantially the same resonant frequencies. C1 is tuned with L1 to aprimary resonance frequency. C2 is tuned with L2 to a secondaryresonance frequency. A series resonant circuit formed by C1 and L1 onthe primary side and a parallel resonant circuit formed by C2 and L2 onthe secondary side, coupling a voltage source V1 to a load resistor RL.The value M is determined equal the one shown in FIG. 1b . The secondaryquality (Qs) is determined by Qs=RL·ω·C2. As greater RL is, the greateris Qs.

The transmission characteristics of the equivalent circuits of FIGS. 1band 1c are identical, they differ only in their resonant circuittopologies. These resonant circuit topologies can be combinedarbitrarily. E.g., the resonant circuit in the source unit and theresonant circuit in the load unit can have the same topology and can beapplied to the described invention as further embodiments. The absolutevalue of the transfer function I2 versus I1 according to FIG. 1brespectively V2 versus V1 according to FIG. 1c is shown in FIG. 1d . Thephase response of the input impedance respectively the input admittanceof the wireless power transmission link is shown in FIGS. 1e . . . 1 g.

Curve 1 shows the transfer characteristics for energy coupling (k·Q)smaller than one. The amount of the transfer function reaches a smallmaximum value and the phase function has a continuous slope during thephase transition from +90 to −90 degrees (see FIGS. 1d and 1e ). Thischaracteristic is called undercritical coupled.

Curve 2 shows the transfer characteristics for energy coupling (k·Q)approximately equal to one. The amount of the transfer function reachesthe maximum value without dip and the phase function has a continuousslope with a minor zero gradient range (without changing the pitch sign)within the phase transition from +90 to −90 degrees (see FIGS. 1d and 1f). This characteristic is called critically coupled and determines theupper energy coupling boundary value.

Curve 3 shows the transfer characteristics for energy coupling (k·Q)greater than one. The amount of the transfer function reaches twomaximum values and the phase function undergoes a sign change in theslope gradient within the phase transition from +90 to −90 degrees (seeFIGS. 1d and 1g ). This characteristic is called overcritical coupledand is avoided in the wireless power transmission link in accordancewith a first aspect of the invention. In the topology according to FIG.1b , it is easy to see that the energy coupling (k·Q) increasesresponsive to a decrease in RL. This corresponds to a load increasewhich acts toward the power source.

Load coupling by means of a series resonant circuit in the load unit istherefore suitable for a voltage source power supply operation (adesired constant output voltage on the load unit side requires a lowsource impedance). In fact, a greater load (smaller RL) couples itselftighter to the source unit due to the greater energy coupling. Similaris true for FIG. 1c . Here, the energy coupling (k·Q) increases when RLincreases. This corresponds to a load increase which acts toward thepower source.

Load coupling by means of a parallel resonant circuit in the load unitis therefore suitable for a current source power supply operation (adesired constant output current on the load unit side requires a highsource impedance). In fact, a greater load (larger RL) couples itselftighter to the source unit due to the greater energy coupling.Regardless of the resonant circuit topology in the source- and load unitthe source unit on the primary side sees a transformed load resistancethat damps the resonant network. The resulting overall quality (Qtot) ofthe wireless transmission path determines the efficiency and frequencyselection of the source unit. A high Qtot represents a high frequencyselection, large reactive power with respect to output power andtherefore a rather lower efficiency. A low Qtot represents a lowfrequency selection, a small reactive power with respect to output powerand therefore a rather higher efficiency.

FIG. 2 depicts a block diagram of a source unit 200 and a load unit 250in accordance with a first embodiment of the current invention. Anenergy source 201 supplies an amplifier 202. 202 drives a parallelresonant circuit formed by L1 and C1. L1 is the primary coil of thewireless power transmission link. A zero- or sign detector 203 generatesan output signal indicative of the zero crossing of the resonancevoltage.

A zero- or sign detector 203 a using a current sensor 203 b generates asignal indicative of the zero crossing of the resonance current. 203 and203 a are substantially identical. The output signal of 203 a supplies afunction controller 206. An output signal of 203 is coupled to a datademodulator 205. 205 operates substantially phase- or frequencysensitive and couples demodulated data from the load unit (FSKLdata) to206. The output signal from 203 is also coupled to block 206 and to aperiod processor 204.

204 analyzes input signals time- or frequency sensitive and generates inrelation to a predetermined comparison value, an output signal (OCT),which in turn serves 206 as an input signal. An output signal of 206controls the frequency of a generator 207, which drives 202. For thatpurpose comprises 206 a phase comparator with a subsequent low passfilter.

The phase comparator controls the phase difference of the output signals203 and 203 a by means of block 207 to zero (resonance frequency). Forexample, a phase difference at the output of the phase comparator inside206 increases or decreases the control voltage, which controls 207. Thisincreases or decreases the frequency of the output signal of 207 untilthe phase difference reaches again zero (resonance frequency). Anotheroutput of 206 controls the output voltage or the output current of block201. Consequently, the resonant circuit energy respectively the radiatedpower of the wireless transmission link is controlled.

The secondary coil L2 of the wireless transmission link forms togetherwith C2 a series resonant circuit, which is tuned to fsoll(corresponding to the resonant frequency of L1 and C1).

A rectifier 251 rectifies the resonance voltage and couples it via a lowpass filter comprising Lf, Cf to a switch SW. When SW is closed, theload resistance RL is coupled through a storage inductance Ls to thecharging capacitor Cf. If SW is open and a current flows in Ls, RL isnot coupled with Cf and the diode D forming a freewheel circuit.

The configuration SW, D, Ls, Cs and RL corresponds to a voltagestep-down converter. A step-down controller 255 measures the outputvoltage across RL and generates at its output essentially a pulse widthsignal (PWM signal) to control SW. For this purpose 255 comprising atleast: a voltage reference, a comparator, a step-down loop filter and aPWM generator (not shown). A zero detector 252 signals a sign change inthe resonance voltage.

Alternatively, in a further embodiment, the resonance current or theresonance voltage in the resonant circuit L2, C2 is detected (notshown). Exact zero crossing detection is not absolutely necessary, it isessential to obtain accurate phase signals of the oscillating circuitperiod (or half period). The output signal from 252 serves 255 (thecontained PWM generator), an analog/digital converter 256 and a datamodulator 253 the synchronization signal. Additionally, the outputsignal from processor 252 is coupled to a period processor 254.

254, corresponding to 204, analyzes the input signal time- or frequencysensitive and generates in relation to a predetermined comparison value,an output signal (OCL) that controls block 255. One or more outputvoltages (e.g. the voltage across Cf and/or RL) are in one or moreblocks 256 analog to digital converted and then in 253 compared versus areference value (feedback versus reference comparison). A resultingcontrol difference value is converted in 253 to an appropriatetransmission format and along with other data, which are generatedand/or stored in 253, coupled as FSKLdata to the ModSW. ModSW modulatesby means of Cm the load of 250. FSKLdata is transferred in this mannerto 200. Alternatively, as another embodiment, ModSW controls C2 by meansof Cm (not shown).

The output voltage control or regulation of the current invention can becarried out in various modes. The modes described below are individuallyand/or jointly active at different times as a combination in variousembodiments. 255 signals the selected mode over the connection mode toblock 253. This mode selection signal controls 253, so that the relevantcontrol data (control difference value) of the respective operation modeare transmitted.

In one mode, SW remains for the output voltage regulation always closedor 250 does not include a voltage step down converter (RL is directlyconnected to Cf). Thus, 256 detects the output voltage across RL. Thecontrol difference value (feedback to reference value comparison, i.e.comparison with a first voltage reference value) in FSKLdata correspondsto the output voltage deviation from a desired output voltage across RL.This control difference value is serially transmitted via the wirelesscoupling link to block 205. 206 uses this data in a standard controlfunction (I, PI or PID) and controls block 201. In this manner thecontrol loop is closed from RL over FSKLdata toward block 201.

In another mode, 250 comprises only SW but no voltage step downconverter. The voltage regulation in 250 is implemented by simplyinterrupting the energy flow (opening SW) in 250 and/or is implementedas described above over the closed loop via the control difference valuein FSKLdata and block 206 in 200.

In another mode, SW operates as a PWM controller in the step-downconverter for the output voltage regulation in 250. 256 detects thevoltage across Cf. The control difference value (feedback to referencevalue comparison, i.e. comparison with a second voltage reference value)in FSKLdata corresponds to the voltage deviation from a desired voltageacross Cf. This control difference value is transmitted together with aduty cycle information of SW (its PWM signal or a digital signalcorresponding to the PWM duty cycle of SW) serially over the wirelesscoupling link to block 205.

206 uses this data in a standard control function (I, PI or PID) andcontrols block 201. In this manner the control loop is closed from RLover FSKLdata toward block 201. This permits to optimize the efficiencyof the wireless power transmission link and yet ensures the operation ofseveral load units 250 on one source unit 200. Additionally, the fastloop response (shortest loop) in the output voltage regulation via SWresults in the best dynamic properties.

A larger output voltage control range is obtained when the voltagestep-down converter is combined with a voltage step-up converter. Thisis implemented as a further embodiment of the current invention and isshown in FIG. 2 with the dashed components. For this purpose, a diode Dsis coupled in series with the switch SW, and a second switch SWs placedin parallel to the diode D.

Advantageously, D and SWs are combined. In the case 250 operates as avoltage step-down converter (voltage across Cf is higher than thedesired output voltage across RL), 255 act as described above. SWdefines by its PWM the output voltage and SWs can be left open or SWscloses as a slave according to the status of D.

Operates 250 as a voltage step-up converter (voltage across Cf is lowerthan the desired output voltage across RL), SW remains closed over 255.Ds prevents the degradation of the higher voltage across RL toward thelower voltage across Cf. SWs operates with a PWM, which determines overits duty cycle the output voltage across RL. The control differencevalue is taken as in the voltage step-down converter from the voltagevalue across Cf. This control difference value is transmitted togetherwith a duty cycle information of SWs (its PWM signal or a digital signalcorresponding to the PWM duty cycle of SWs) serially over the wirelesscoupling link to block 205.

206 uses this data in a standard control function (I, PI or PID) andcontrols block 201. In this manner the control loop is closed from RLover FSKLdata toward block 201. Advantageously, this combined step-uprespectively step-down configuration stabilizes the output voltage forboth lower and higher received voltages on Cf.

It is also readily apparent that one or more 250 comprise one or moreoutput voltages.

In another mode, the control difference value is derived from aplurality of output voltages, which are sampled in a time multiplexmanner and/or be added through resistors. This control difference valueis transmitted along with a duty cycle information of one or moreswitches SW (their PWM signals or a digital signal corresponding to thePWM duty ratios of a plurality of switches) serially over the wirelesscoupling link to block 205. 206 uses this data in a standard controlfunction (I, PI or PID) and controls block 201. In this manner thecontrol loop is closed from RL over FSKLdata toward block 201. If aplurality of 250 operate with one 200, 206 combines multiple controldifference values of a plurality of received FSKLdata of coupled 250 andcontrols 201. In this way, the total radiated power of 200 is adjustedto the overall power level required to supply a plurality of 250. Thefine output voltage regulation within the individual 250 is controlledby SW and/or SWs.

It is also possible to operate one or more 250 with continuously closedSW together with one or more 250, which use SW and/or SWs as outputvoltage regulator on one 200.

These output voltage control or regulation methods respectively theirdifferent modes correspond to the normal operating mode (continuousoperation), while OCL is not active. For this case, the periodcomparison versus a predetermined reference value generates inside 250 afirst result, which resets OCL to OFF (see FIG. 4). This corresponds tothe operating condition during the energy coupling conditioncharacteristic for curve 1 in FIGS. 1c and 1 d.

A further increase in the energy coupling (higher coupling k and/orlarger load) results in critical- and finally overcritical couplingcondition. The input impedance develops a plurality of differentresonant frequencies (every zero crossing of the phase line of the inputimpedance). This causes in the control loop 202, 203, 203 a, 203 b, 206and 207 a phase jitter around to the resonance frequency at criticalcoupling condition, which in excess leads to a superposition of twocurrent or voltage curves at overcritical coupling condition. This stateis defined here as over coupling and defines an unwanted operationstatus. The state change from critical toward overcritical couplingcondition arises fluently in practice.

In practice, it is sufficient to detect the state of overcriticalcoupling explicitly for the energy coupling control of the wirelesspower transmission link. Depending on the implementation of thefunctions described in the following, 204 respectively 254 respondsfaster (even at critical coupling condition) or responds slower (notuntil overcritical coupling condition). 254 in 250 compares the halfperiod of the detected zero-crossings versus a predetermined comparisonvalue, which is a little less than the half period of the operationfrequency in undercritical coupling condition. This comparison nowyields a second result, which changes the state of OCL to ON (highpotential).

In one embodiment, the period processor compares multiple of detectedhalf periods versus almost the same value corresponding to the samemultiple of half periods of the operating frequency in undercriticalcoupling condition. The response of the coupling detector 254 increasesaccordingly.

The ON in OCL forces 254 to generate an output signal, which opens SW aslong as OCL is ON. The function of the voltage step-down converter isdisabled and RL is decoupled from Cf. OCL remains ON even if the outputof 252 indicating normal energy coupling conditions (detected halfperiod is greater than the predetermined comparison value).

As a consequence, the lower load reduces the energy coupling. After anarbitrary time (TL) where 250 remains unloaded (RL is decoupled fromCf), the output of 254 changes independently (e.g. by means of animplemented one shot circuit or timer), OCL goes to OFF and RL restorescoupling with Cf by closing SW (see FIG. 4). The step-down converterrestores normal operation.

When k and RL remain unchanged, the overcritical coupling conditionreestablishes as soon as the system reaches the steady state conditionin Lf and Cf. SW reopens via OCL for the time interval TL, then closesagain etc.

Consequently, the coupling status of RL with Cf alters continuously. Theaverage value of the effective RL appearing in the energy couplingreaches in this manner the value which is substantially equal to theboundary value of RL defining the critical coupling condition. Thisrepresents a limitation of the energy coupling to a maximum of one.Since 254 is responsive to one or a few half-periods and TL is muchlonger, the operation status during which the wireless powertransmission link operates in the overcritical coupled condition reducesto a minimum or is prevented completely. Thereby, the entire powertransmission link operates in the source- and load unit substantiallycontinuously under resonant condition, i.e. at the resonance frequency.This resonance frequency is determined by 200 and 250.

Advantageously this results in the smallest possible bandwidth.Additionally, the entire power transmission link operates always undermatched conditions because always only the real part of the entiretransmission path changes. A resonance frequency detuning is alwaysreadjusted in 200. The effective real load resistance of 200 isdetermined by RL, the turn ratio of L1 and L2, the window function (dutycycle) of the signal OCL and the energy coupling (k·Q). Thus, thewireless power transmission link directly corresponds to an RL(eventually transformed by L1/L2) contacted at 200 and thus correspondsto a wired coupling. By choosing TL and selecting the components of thefilter Lf and Cf, the ON/OFF timing of OCL can be designed flexibly. Anabrupt coupling respectively abrupt decoupling of RL has no adverseimpact on 200, since Lf and Cf act as a filter, which smoothing anyappearing load transients on 200.

In 200 occurs something similar. If an overcoupling condition isdetected in 204, OCT sets to ON, 206 disables 201 or reduces the outputpower of 201 to a minimum value for a time interval TT. The timeresponse of OCT is delayed by TD versus OCL. After the time interval TThas elapsed, the output of 204 changes independently (e.g. by means ofan implemented one shot circuit or timer), OCT goes to OFF and 201 isagain enabled or restored to normal output power (see FIG. 4). Ifovercoupling condition is redetected, the process repeats: 201 againdisables delayed by TD or reduces the output power and automaticallyreestablishes normal operation after the interval TT. In this manner asafe operation and a soft start at 200 is guaranteed wheneverovercoupling condition occurs.

In one embodiment of the current invention, 204 is featured with adelayed time response versus the time response of 254 concerning thestart of TD, for an appearing overcoupling condition.

In another embodiment, 204 and 254 are featured with an identicalresponse time (same or similar implementation), then the required delayin the start of TD is implemented in 206 as a delay on the event OCT.Operationally, both versions are identical. TD ensures that the energycoupling control is always first executed at 250. Advantageously, other250, which are coupled to 200 remain continuously supplied with energydue to the continuous operation of 200.

TT within 200 and TL within 250 need not to be equal. The redundancy ofthe double implementation of 208 within 200 and 250 limit, in any case,the energy coupling (k·Q) substantially to the value one. This comprisesalso a possible malfunction in 250: Then, the other block 208 reacts assoon as overcoupling condition has established.

A source unit can also radiate energy in a predetermined frequencyrange. Further, a load unit can drive a load as a regulated currentsource. This is illustrated in the block diagram in FIG. 3. The sourceunit 300 comprises a large signal Voltage Controlled Oscillator (VCO)301, which is powered with the necessary operating voltage by acontrolled power source 302.

The term large signal VCO stands for an oscillator whose active element304 (amplifier) operates mainly as a switch. Further, the frequency ofthis oscillator is controlled by a current or a voltage. Advantageously,the frequency controller comprises at least one coupling switch, whichcouples at least one inductor or at least one capacitor via a variablecoupling interval to the resonant circuit, wherein the coupling intervalis smaller than a period of the resonant circuit.

The inverting amplifier 304 drives a series resonant circuit comprisingthe electrically controlled capacitor C1 and the primary coil L1 of thewireless power transmission link. An inverter 303 closes a feedback loopto the input of 304 and provides a continuous current- and voltageoscillation in the resonant circuit (positive feedback).

In a further embodiment of the present invention, the large signal VCOis equipped with an H-bridge or push-pull stage and is implemented witha parallel resonant circuit (not shown in FIG. 3). The functions of allthe properties described (Phase Locked Loop (PLL) operation and controlloops) remain the same as for the large signal VCO driving the seriesresonant circuit.

The oscillator signal (output signal first of 303) is compared in aphase comparator 305 versus a reference value (fsoll). fsoll isgenerated in a frequency synthesizer 306 using a fundamental frequency(fref). The output signal from 305 is filtered by a loop filter 307 andsubsequently coupled as a control factor to the variable capacitor C1.When 301 does not oscillate at fsoll, 305 generates an error outputsignal, which retunes C1 after filtering with 307 until first becomesequal to fsoll. In this manner 300 emits electromagnetic energy, whichis regulated in the frequency at fsoll. Any kind of frequency detuningsuch as by component tolerances, component aging or changes in the loadunit 350 are corrected within a few dozen oscillator periods. Thecoupling detector 308 comprises the period processor 309, because thezero-crossing- or sign detector essentially corresponds to 303. Anoutput signal of 305 is coupled to a data demodulator 311.

Alternatively, in another embodiment, 311 uses first as its inputsignal. 311 operates substantially phase or frequency sensitive andcouples demodulated data from the load unit (FSKLdata) to a functioncontroller 310. The output signal of 303 is coupled to a periodprocessor 309. 309 analyzes the input signals time- or frequencysensitive and generates in relation to a predetermined comparison value,an output signal (OCT), which in turn serves 310 as an input signal. Anoutput signal of 310 controls 306 and defines in this manner fsoll.

For this purpose comprises 306 mainly a programmable frequency divider.In one embodiment of the actual invention the frequency divider within306 changes its divisor as a function of time to form an arbitraryfrequency spectrum (e.g. approximated rectangular spectrum or Sin(x)/x,etc.) in fsoll and accordingly in the radiated field. Another output of310 controls the output voltage or the output current of 302.

The secondary coil L2 of the wireless power transmission link formstogether with C2 a parallel resonant circuit, which is tuned to fsoll(corresponding to the resonance frequency of C1 and L1). If fsoll ischanged over time, the resonance frequency of L2 and C2 is selected sothat it substantially matches the center of the frequency range, whichin turn is radiated by 300.

A rectifier 351 a, 351 b rectifies the resonance voltage and couples itthrough a low pass filter LF and a diode Dp to a filter capacitor Cf andthe load RL. Parallel to 351 a and 351 b are each a switch SWa and SWbconnected. Advantageously, 351 a and SWa, respectively, and 351 b andSWb are integrated into one component. If SWa and SWb are open, RL iscoupled via 351 a during one half resonant circuit period and via 351 bduring the other half resonant circuit period over Lf to the resonantcircuit L2, C2.

If SWa and SWb are closed at the same time, the resonant circuit isshunted by SWa and SWb, Dp blocks and RL is no longer coupled with theresonant circuit L2, C2. The configuration SWa, SWb, C2, L2, Lf and Dpoperate as a shunt regulator for RL. An analog- to digital converter 356measures by means of a current sensor 356 a the load current through RL.355 generates responsive to the output signal of 356 a pulse widthsignal (PWM signal) and controls SWa and SWb. For this purpose 355comprises at least: a voltage or current reference, a comparator, a loopfilter and a PWM generator (not shown). A zero-crossing detector 352signals a sign change in the resonance voltage.

Alternatively, in a further embodiment, the resonance current or theresonance voltage in the resonant circuit L2, C2 is detected (notshown). The output signal from 352 synchronizes block 355 (the containedPWM generator), block 356 and the data modulator 353. In addition, theoutput of 352 is also coupled to a period processor 354. 354corresponding to 309 analyzes the input signals time- or frequencysensitive and generates in relation to a predetermined comparison value,an output signal (OCL) that controls block 355.

The output value of 356 is compared in 353 with a reference value(feedback versus reference comparison). A resulting control differencevalue is converted in 353 to an appropriate transmission format andalong with other data, which are generated and/or stored in 353 coupledas FSKLdata to ModSWa and ModSWb. ModSWa and ModSWb modulate Cm with theload of 350. FSKLdata is transmitted in this manner to 300.

Alternatively, in a further embodiment, Cm is coupled at the tapping ofL2 via a ModSW to ground.

The output current control or regulation of the current invention can becarried out in various modes. The modes described below are individuallyand/or jointly active at different times as a combination in variousembodiments. 355 signals the selected mode over the connection mode to353. This mode selection signal controls 353, so that the relevantcontrol data (control difference value) of the respective operation modeare transmitted.

In one mode, SWa and SWb remain for the output current regulation open.Thus, 356 detects the output current in RL. The control difference value(feedback to reference value comparison, i.e. comparison with a firstreference value) in FSKLdata corresponds to the output current deviationfrom a desired output current through RL. This control difference valueis serially transmitted via the wireless coupling link to block 311. 310uses this data in a standard control function (I, PI or PID) andcontrols block 302. In this manner the control loop is closed from RLover FSKLdata toward block 302.

In another mode, SWa and SWb operate as a PWM controller for the outputcurrent regulation in 350. 356 detects the current through RL. Thecontrol difference value (feedback to reference value comparison, i.e.comparison with a second reference value) in FSKLdata corresponds to thecurrent deviation from a desired current through RL. This controldifference value is transmitted together with a duty cycle informationof the switches SWa respectively SWb (its PWM signal or a digitalsignals corresponding to the PWM duty cycle of SWa respectively SWb)serially over the wireless coupling link to 311. 310 uses this data in astandard control function (I, PI or PID) and controls block 302. In thismanner the control loop is closed from RL over FSKLdata toward block302.

This permits to optimize the efficiency of the wireless powertransmission link and yet ensures the operation of several load units350 at one source unit 300. Additionally, the fast loop response(shortest loop) in the output current regulation by means of SWarespectively SWb results in the best dynamic properties.

It is also readily apparent that one or more 350 may comprise one ormore output currents.

In another mode, the control difference value is derived from aplurality of output currents, which are sampled in a time multiplexedmanner and/or are simply added. This control difference value istransmitted along with a duty cycle information of one or more switchesSWa respectively SWb (their PWM signals or a digital signalcorresponding to the PWM duty ratios of a plurality of switches)serially over the wireless coupling link to block 311. 310 uses thisdata in a standard control function (I, PI or PID) and controls block302. In this manner the control loop is closed from RL over FSKLdatatoward block 302.

If a plurality of 350 operate with one 300, 310 combines multiplecontrol difference values of a plurality of received FSKLdata of coupled350 and controls 302. In this way, the total radiated power of 300 isadjusted to the overall power level required to supply a plurality of350. The fine output current regulation within the individual 350 iscontrolled by SWa respectively SWb.

It is also possible to operate one or more 350 with continuously openSWa respectively SWb together with one or more 350, which use SWa andSWb as output current regulator on one 200.

These output current control methods respectively their different modescorrespond to the normal operating mode (continuous operation), whileOCL is not active. For this case, the period comparison versus apredetermined reference value generates inside 350 a first result, whichresets OCL to OFF (see FIG. 4). This corresponds to the operatingcondition during the energy coupling condition characteristic for curve1 in FIGS. 1c and 1d . A further increase in the energy coupling (highercoupling k and/or larger load) results in critical and finallyovercritical coupling condition. The input impedance develops aplurality of different resonant frequencies (every zero crossing of thephase line of the input impedance). This causes in the control loop 301phase jitter around the resonance frequency at critical couplingcondition, which in excess leads to a superposition of two current orvoltage curves at overcritical coupling condition. This state is definedhere as over coupling and defines an unwanted operation status. Thestate change from critical-toward overcritical coupling condition arisesfluently in practice. In practice, it is sufficient to detect the stateof overcritical coupling explicitly for the energy coupling control ofwireless power transmission link. Depending on the implementation of thefunctions described in the following, 309 respectively 354 respondsfaster (even at critical coupling condition) or responds slower (notuntil overcritical coupling condition). 354 in 350 compares the halfperiod of the detected zero-crossings versus a predetermined comparisonvalue, which is a little less than the half period of the operatingfrequency in undercritical coupling condition. This comparison nowyields a second result, which changes the state of OCL to ON (highpotential).

In one embodiment, the period processor compares multiple of detectedhalf periods versus almost the same value corresponding to the samemultiple of half periods of the operation frequency in undercriticalcoupling condition. The response of the coupling detector 354 increasesaccordingly. The ON in OCL forces 354 to generate an output signal,which closes SWa and SWb as long as OCL is ON. Consequently, RL isdecoupled from C2, L2 by Dp. OCL remains ON even if the output of 352indicating normal energy coupling conditions (detected half period isgreater than the predetermined comparison value).

As a consequence, the lower load reduces the energy coupling. After anarbitrary time (TL) where 350 remains unloaded (RL is decoupled fromC2), the output of 354 changes independently (e.g. by means of animplemented one-shot circuit or timer), OCL goes to OFF and RL restorescoupling with C2, L2 by opening SWa and SWb (see FIG. 4). The rectifier351 a, 351 b restores normal operation. When k and RL remain unchanged,the overcritical coupling condition reestablishes as soon as the systemreaches the steady state condition in Lf and Cf. SWa and SWb close againvia OCL for the time interval TL, then open again, etc. Consequently,the coupling status of RL with C2, L2 alters continuously. The averagevalue of the effective RL appearing in the energy coupling reaches inthis manner the value which is substantially equal to the boundary valueof RL defining the critical coupling condition. This represents alimitation of the energy coupling to a maximum of one.

Since 354 is responsive to one or a few half-periods and TL is muchlonger, the operation status during which the wireless powertransmission link operates in the overcritical coupled condition reducesto a minimum or is prevented completely. Thereby, the entire powertransmission link operates in the source- and load unit substantiallycontinuously under resonant condition, i.e. at the resonance frequency.This resonance frequency is determined by 300 and 350. Advantageouslythis results in the smallest possible bandwidth. Additionally, theentire power transmission link operates always under matched conditionsbecause always only the real part of the transmission path changes. Aresonance frequency detuning is always readjusted in 300. The effectivereal load resistance of 300 is determined by RL, the turn ratio of L1and L2, the window function (duty cycle) of the signal OCL and theenergy coupling (k·Q).

Thus, the wireless power transmission link directly corresponds to an RL(eventually transformed by L1/L2) contacted at 300 and thus correspondsto a wired coupling. By choosing TL and selecting the components of thefilter Lf and Cf, the ON/OFF timing of OCL can be designed flexibly. Anabrupt coupling respectively abrupt decoupling of RL has no adverseimpact on 300, since Lf and Cf act as a filter, which smoothing anyappearing load transients on 300.

In 300 occurs something similar. If an overcoupling condition isdetected in 309, OCT sets to ON, 310 disables 302 or reduces the outputpower of 302 to a minimum value for a time interval TT. 350 is no longersupplied with energy. The time response of OCT is delayed by TD versusOCL. After the time interval TT has elapsed, the output 309 changesindependently (e. g. by means of an implemented one shot circuit ortimer), OCT goes to OFF and 302 is again enabled or restored to normaloutput power (see FIG. 4). If overcoupling condition is redetected, theprocess repeats: 302 again disables delayed by TD or reduces the outputpower and automatically reestablishes normal operation after theinterval TT. In this manner, a safe operation and a soft start at 300 isguaranteed whenever overcoupling condition occurs.

In one embodiment of the current invention, 309 is featured with adelayed time response versus the time response of 354 concerning thestart of TD, for an appearing overcoupling condition.

In another embodiment, 309 and 354 are featured with an identicalresponse time (same or similar implementation), then the required delayin the start of TD is implemented in 310 as a delay on the event OCT.Operationally, both versions are identical. TD ensures that the energycoupling control is always executed at 350. Advantageously, other 350,which are coupled to 300 remain continuously supplied with energy due tothe continuous operation of 300.

TT within 300 and TL within 350 need not to be equal. The redundancy ofthe double implementation of 308 within 300 and 350 limit, in any case,the energy coupling (k·Q) substantially to the value one. This comprisesalso a possible malfunction in 350: Then, the other block 308 reacts assoon as overcoupling condition has established.

In one embodiment of the current invention, 310 changes the resonancefrequency in 300 responsive to 306 and 305. This represents anothervariant to reduce the energy coupling. In this manner, advantageously,the coupling k and/or the quality factor Q of 350 and above all theamplitude of 301 change. This may be sufficient to inhibit theovercoupling condition. Thus, 300 features an additional degree offreedom versus 200 by arbitrarily controlling the resonance frequency.This resonance frequency change occurs alone or combined with a controlof 302.

Advantageously, the described methods of the different modes of 200,250, 300 and 350 of FIGS. 2 and 3 serve greatest flexibility. Each 250and 350 may determine itself the method by which the output voltagerespectively the output current should be regulated. The controldifference value, which is transmitted in FSKLdata describes a measureas to whether more or less power over the wireless transmission linkfrom 200 respectively 300 is to be pushed to 250 respectively 350.

In another embodiment of the current invention are, in the event ofovercoupling condition, only individual partial loads, i.e. a number ofoutput voltages of a plurality of output voltages in 250 respectively anumber of output currents of a plurality of output currents in 350,successively decoupled via a plurality of switches (SW), or diodes (DP)within 250 respectively 350. In this manner, other partial loads of 250respectively 350 remain continuously supplied with energy.

In general, the energy coupling limitation of the current inventionlimits the load energy or power. Depending on the implementation of thefunctions and operating modes described, an output voltage across RL andrespectively an output current through RL may more or less drop. Thanksto the existing energy storages (Ls, Cs within 250 respectively Lf, Cfwithin 350), the output voltage respectively the output current mightnot interrupt necessarily. All modes of the described methods mayinteroperate with each other and the pairing of any one or more 250respectively 350 with one 200 respectively 300 can be permutedarbitrarily. Advantageously, TL respectively TT is elected by factorsgreater than the lock time of the PLL loop 202, 203, 203 a, 203 b, 206,and 207 respectively 301, 305 and 307.

FIG. 5a shows a typical flexible implementation according to oneembodiment of the current invention: A transmitter and/or receivercoupling coil (LNFC) is used in two different frequency ranges at thesame time and/or at different times in two different main applications.

The first main application is an RFID or Near Field Communication (NFC)by inductively coupled inductors and operates in a first frequency rangeabove 1 MHz (e.g. 6.78 MHz or 13.56 MHz). The capacitors CNFC1 and CNFC2tune LNFC to the first frequency range and/or match LNFC to the input oroutput of the processing unit 500 of the first main application. Inaddition, CNFC1 and CNFC2 decouple an operating status of the first mainapplication from an operating status such as transmission or receptionof the second main application.

Additionally, in a further embodiment of the invention, switches oramplitude limitation means (e.g. diodes) are implemented to protect theinput or output stage within 500 from overvoltage as depicted in FIGS.5b-5h . E.g. limiting diodes (D) connected to a reference potential areused for that purpose. The switches (SW1, SW2) are in series (SW1)and/or implemented as switches (SW2) to a reference potential.

The second main application is a wireless power transmission, andoperates in a second frequency band below 1 MHz (for example, 120 . . .135 kHz). The coils Lx and Ly tune LNFC to the second frequency rangeand/or match LNFC to the input or output of the processing unit200/250/300/350 of the second main application. In addition, Lx and Lydecouple an operating status such as transmission or reception of thesecond main application from an operating status of the first mainapplication. The processing unit 200/300/250/350 corresponds to one ormultiple blocks in FIG. 2 respectively FIG. 3, wherein multiple blockscan be operational at different times. In one embodiment of the actualinvention the second main application operates as a transmitter andemits power by means of LNFC.

In another embodiment of the current invention, the second mainapplication operates as a receiver and receives power by means of LNFC.Advantageously, 250 operates in the second main application as a voltagestep-up converter, since the induced voltage due to the low inductanceof LNFC is relatively small (in the range 4 uH . . . 20 uH) and thedecoupling inductances Lx and Ly are rather large (in the range 30 . . .100 uH).

If a plurality of 250 or 350 operate on one 200 or 300, thecommunication of the individual FSKLdata may be disturbed. For thisreason, the individual FSKLdata are transmitted in periodic- or randomtime division multiplex. As a consequence, ModSW remains open or closedif no FSKLdata are transmitted.

FIG. 6a shows two at different times transmitted FSKLdata1 andFSKLdata2, whereby these data might be generated from the same or twodifferent 250 or 350. During the time intervals Tbusy no furtherFSKLdata can be sent because the channel is busy (the channel busyindicator FSKLdatajam is active). 253 and 353 comprise for this purposea communication detector 601 (see FIG. 6b ), which signals the timeinterval Tbusy. The input signal of block 601 is served by synch. Synchis analyzed within 601 in a time and/or frequency sensitive manner andgenerates at its output a binary signal COMready, which is OFF duringthe time interval Tbusy, otherwise set to ON.

The communication trigger 602 generates a positive output pulse COM in aperiodic or random manner having the length of FSKLdata, wherein thepositive output pulse COM is only generated during Tvalid. This positiveoutput pulse COM signals the communication controller 603 to generate aserial bitstream of FSKLdata (or at least portions thereof) by means ofthe clock COMclk and couples FSKLdata to ModSW. COMclk is derived fromsynch by a frequency divider 604. In this manner, advantageously,FSKLdata are always sent synchronously with respect to the resonantcircuit periods. This simplifies 205 and 311 and provides good receptionsensitivity.

FSKLdata transmits binary information, which is differentially encodedin two phases at a bitrate whose clock is divided from synch. The binaryinformation is formatted in byte length. The communication protocolFSKLdata comprises data formatted in a header (constant bit length), themessage (variable bit length), and a checksum (constant bit length).

The header defines the message portion of the following message and itslength in bytes. E.g. the lower 4 bits of the header defines the messageportion and the upper 4 bits defines the message length (see FIG. 7). Inthis case, one or more message portions can vary in their length (e.g.control variable in FIG. 7f ). The message transmits the messageportion, which was previously signaled in the header. The checksumtransmits a CRC- or hash value calculated over the message portion oralternatively also calculated over the message including the header.

In a first message portion at least one identification number which ispre-stored in 250 or 350 is transmitted (see FIG. 7a ).

In a further embodiment of the current invention, an identificationnumber is generated for a time interval (e.g. duration of energytransmission session).

In another message portion, an identification number is transmitted (seeFIG. 7b ), which characterizes a communication protocol version of themessage (header/message portion encoding according to the pattern asshown in FIG. 7 or similar).

In another message portion, at least one value characterizing power orcharacterizing a power class is transmitted (see FIG. 7c ).

In a further message portion, at least one value of a secondary receiverinductor (L2) and/or the number of windings of L2 is transmitted (seeFIG. 7d ).

In a further message portion, at least one value of a resonancefrequency and/or a reception bandwidth is transmitted (see FIG. 7e ).

In another message portion, at least a control difference value or acontrol trend is transmitted (see FIG. 7f ).

In another message portion, at least a value of a coupling status or anenergy coupling status (OCL flag) and/or a operation status istransmitted (see FIG. 7g ). The OCL flag is a bit that is set by the OCLsignal at 253 respectively 353.

In another message portion, at least a value of a not allowed frequencyrange (i.e. an excluded transmission frequency or bandwidth) istransmitted (see FIG. 7h ).

200 or 300 reads these values and adjusts operation mode conditionsaccording to the received message or message portion.

Thus, in one embodiment of the current invention, at least one or moremessage portions of the received message are processed according to thereceived protocol identification.

In another embodiment, 201 respectively 302 is controlled in the outputpower responsive to the control difference value, and if necessarylimited in the output power responsive to the power class and/or to theinductor L2 and/or to the number of turns of L2. E.g. if a controldifference value reaches a predetermined value, then the radiated poweris limited or reduced until the received control difference value doesno longer indicate the predetermined value. This guarantees overdriveprotection of all 250 and 350.

Additionally, as an option, 201 respectively 302 is enabled respectivelydisabled (continuous energy transmission respectively no energytransmission) or continuously switched between these two operationstates (burst mode) responsive to the received operation status and/orthe received energy coupling status information (OCL flag) in FSKLdata.

In a further embodiment, 306 is controlled in fsoll such that theradiated power at a predetermined resonance frequency or over afrequency bandwidth, under consideration of a not allowed frequency orfrequency range, is in accordance with a predetermined value or at leastapproximates the predetermined value.

In a further embodiment, 310 is controlled such that the emittedelectromagnetic power is disabled or at least the emitted power islimited within a frequency range responsive to a specific identificationnumbers. In this manner, the power flow to individual 250 or 350 iscontrolled, or the power flow is enabled or disabled responsive to anidentification number.

FIG. 8 shows an example of a detailed circuit according to 300 in FIG.3. FIGS. 9a and 9b show signal waveforms, which are referred to in thefurther description. A large signal VCO, here a parallel push-pullcircuit with Q1 and Q2 drives the parallel resonant circuit comprisingL1, C1. DR energizes the center tap of L1 by a controllable power supplySMPS. Advantageously, the voltage across L1 in this resonant circuittopology is independent of the quality of the resonant circuit and isfurther proportional by a constant to the supply voltage at the outputof the SMPS. R8, D3 and D4 switch Q5 always conductive, except duringthe voltage zero crossings of the resonant circuit voltage.

The voltage across R7 clocks the D flip-flop FF1, which operates as afrequency divider. The output signals are delayed by N1 . . . N4 andfurther coupled in an AC manner via CK1, CK2, D1, D2, R5 and R6 to Q6 .. . Q9. Q6 . . . Q9 drive Q1 and Q2 by means of R1 and R2. The outputsignal first of Q5 is also coupled to the input of the phase comparatorPFD of block 801, which compares first versus fsoll. 801 correspondssubstantially to 305 and 307.

The charge pump R9, R10, Q14, Q15, Q16 and Q17 generates in the PLL loopfilter Ci, Cp and Rp, an error voltage corresponding to the phasedifference between first and fsoll. This error voltage generates withthe current in Q18 . . . Q20 a ramp voltage across Cr, that is modifiedby repeating short conducted Q21 to a sawtooth voltage (waveform C).First is formed by NOR and N6 into short time pulses now named as synchthat controls Q21 and synchronizes the FSKLdata demodulation in thecontroller (signal curve synch).

This sawtooth voltage alters the state of N5, as soon as its thresholdvoltage is reached (see FIG. 9b T1, T3, T4, T6 intersections with thedotted line curve C and output D). The following OR 1 and OR2 route thetiming signal D to the corresponding coupling switches Q3 respectivelyQ4 to control the resulting value of capacitor C1 in a large signalmanner. The driving signals of FF1 are inverse to each other and changetheir state at 0, T3, and T6. In this manner, only every second periodof D is routed to E respectively F. The drivers Q10 . . . Q13 switch Q3respectively Q4 accordingly via R3 and R4. As a consequence, across Q3respectively Q4 develops alternately a half sine wave per period (seewaveform VQ3 dashed respectively VQ4). R1 . . . R4 have a low value andprevent transients in the switch control. Q3 and Q4 closing during theinterval T2, T3 (Q4) and the interval T5, T6 (Q3) independently throughtheir internal diodes as soon as C1′ respectively C1″ has discharged.

This phase control of the effective parallel capacitance of C1′ and C1″provides a control range of C1tot=C1+0.5·C1′, if C1′=C1″. Fist is inthis manner always perfectly adjusted to fsoll. Fref is divided by theprogrammable divider synth to obtain fsoll. Delays N1 . . . N4compensate the propagation delay of fist to D and ensure the fullcontrol range in C1tot. One output of FF1 is coupled to the clock inputsof FF2 and FF3. FF2 is set with the rising edge and after reaching thethreshold over CT1 reset. RT1 and CT1 define the time constant(predetermined reference value).

If signal curve v reaches the threshold level of the reset input of FF2,then only a “OFF” is latched to the output of FF3 (see waveform u). Ifthe threshold level of the reset input of FF2 is not reached (becausedrive operates at a too high frequency), then a “ON” is latched to theoutput of FF3. This situation occurs after the second drive period inFIG. 9a , since drive in this example has such a high frequency that vnever reaches the threshold (continuous over-coupling).

D5 ensures the reset of the voltage across CT1. The mono stableflip-flop FF4, RTT and CTT generate a high (ON) in OCT when u remainsset for more than one drive period. After a time interval (TT) haselapsed, the voltage across CTT reaches the threshold, which resets FF4.OCT is coupled to the controller, if necessary, internally in thecontroller additionally delayed, and then used over PBUS to control theSMPS. A signal responsive to OCT is signaling the energy coupling statusvisually by means of LED D7 (over coupling indicator).

The controller comprises the divider synth and also the FSKLdatademodulator. In this way, the frequency deviation is measured by a cyclecounter over a temporal profile. The received FSKLdata control the SMPSand/or synth, and optionally further optical indicators (not shown inFIG. 8). For this purpose, the controller comprises a microcontrollerwith RAM and ROM memory and/or a PLD/FPGA and/or ASIC components. Theseimplementation methods are known to a person skilled in the art.

FIG. 10 shows the detailed circuit of a receiver 250 of a wireless powertransmission system according to FIG. 2. L2 form together with C2 aseries resonant circuit for receiving the electromagnetic field lines.GL rectifies the received signal and couples it after filtering with Lfand Cf to SW within 1000. This block corresponds to a voltage step-downconverter and represents a typical implementation of LM5116 by anexcerpt of its data sheet. D is implemented as a switch and LS, Cs arestorage elements. The exact detail function of LM5116 and its componentsshown here is referred to the data sheet. The operation mode of LM5116is carried out in diode emulation mode, or alternatively in thesynchronous mode. Rfb1 and Rfb2 couple the output voltage across RL in avoltage divider manner to the feedback input FB. The switching frequencyof the PWM generator inside LM5116 is here directly synchronized withthe electromagnetic field. For this purpose, R8, D3 and D4 keep Q5conducting, except during the minima of the rectified resonant circuitvoltage. N6 and NOR form the voltage across R7 into short pulses, whichcontrol over Csync and RT the RT input of LM5116. This synchronizes theinternal PWM and switching frequency of 1000 with the receivedelectromagnetic alternating field received at L2, C2. Advantageously,1000 discharges the resonant circuit L2, C2 (filtered by LF, Cf) on aresonant circuit half-period base and the energy charging respectivelythe energy discharge of the resonant circuit L2, C2 and its loadsubstantially coincides. This reduces the required capacitor size andcurrents in Cf versus the unsynchronized case with the same filterproperties. As a consequence, the cost reduces and the reliabilityincreases.

In addition, the NOR output (synch) is coupled to the synchronizationinput of the Controller/AD. The voltage across R7 is frequency dividedin FF1 and thereafter analyzed in 1001 whether overcoupling conditionhas developed or not. 1001 corresponds to 802, and hence the waveformsof FIG. 9 are valid. The output signal of 1001 (OCL) is inverted by Nenand coupled to the EN input of LM5116. If overcoupling condition isdetected (OCL is at high level), thus SW opens and the voltage downconverter interrupts the energy flow from Cf to RL.

OCL is processed together with other values that shall be transmittedwithin the Controller. Here refer to OCT and Controller in FIG. 8. TheController shown in FIG. 8 establishes an additional time response delayfor the operational behavior with the load unit in FIG. 10. By means ofthis delay, the resulting OCT appears always after OCL (see FIG. 4). Thetiming of OCT and OCL in the overall wireless power transmissionprocedure is indicated in FIG. 4.

The timing shown in FIG. 9a represents a quantitative illustration of1001.

In another embodiment of the current invention, 1001 is eliminated inFIG. 10. Here, the signal OCL is directly determined in theController/AD by counting fref clock cycles within one synch period. Forthis case, the Controller according to FIG. 8 can be featured with asmaller or no additional delay.

In FIG. 10, the output voltage across RL, and the voltage across Cf iscoupled by means of an additional voltage divider RF1 and RF2 to theController/AD. The existing analog-digital converter (AD) withinController converts these input values into a digital format.

These data are coupled in the above described manner (protocol formationand redundancy addition), and after adding other values (see FIG. 7) asFSKLdata via the drivers and R4 to ModSW. ModSW sets Cm while numbers ofresonant circuit periods corresponding to the bit rate, to ground, andmodifies in this manner the capacitance of C2.

The bit rate is obtained from down divided synch in the Controller/AD ofFIG. 6b . In the case the Controller/AD is implemented as sequentialprogram (software), then the implementation of FIG. 6b is slightlydifferent. It is important that the data rate (bit rate) is synchronizedwith synch and a possible data transfer conflict (FSKLdatajam) isavoided. For this purpose, the Controller/AD detects period- orfrequency changes of the signal synch and any received FSKLdata of otherload units by means of reference frequency.

Alternatively, in an embodiment of the present invention any otherFSKLdata are demodulated (e.g. at least a portion of FIG. 7) and shownto the user. The Controller/AD requires beside data processing meansinternal ND converters, RAM and ROM memory, and their coupling means(MBUS) to the controller. The processing frequency (clock) is fref oralternatively multiple factors of synch. R6 and C3 eliminate or reducevoltage transients at the output of GL (snubber). The resulting energyis dissipated across a resistor, or alternatively, as shown here, usedas an additional voltage V+(e.g. for the Controller/AD). For thispurpose D2 charges C4 and D1 limits V+. Alternatively, a coil is coupledin series to D2 (not shown). In this manner, the load unit comprisesalways a minor base load. Further, this base load is also determined byRuv1, Ruv2, Rf1, Rf 2 and the base load of LM5116.

It is obvious for a person skilled in the art that various describedmethods and/or modes of FIGS. 2, 3, 8 and 10 comprise or include digitalsignal processing. This is implemented in a discrete or integratedmanner, as a programmable logic device (PLD, FPGA), and/or as piece ofhardware or software within a microcontroller. Furthermore variousnumbers of sub-blocks within the FIGS. 2, 3, 8 and 10 can be integratedin integrated circuits (ASICs).

In one embodiment of the current invention, synch is served as areference frequency for other communication means, such as e.g. NFC,RFID, Bluetooth, WLAN, UWB, etc. (see output synchout in FIG. 10). Forthis purpose synch is multiplied by means of a PLL to an integer number,and thereafter used to control a data rate (bit rate) or to define andsynchronize the transmission frequency and/or a communication timewindow (communication time slot) of a time multiplex transmission and/orto control a random code sequence of a code division multiplex system(frequency hopping FH- or a direct sequence method).

1-15. (canceled)
 16. A wireless power receiver comprising: at least onefirst capacitor and at least one first inductor forming at least onefirst resonant circuit in said wireless power receiver for receivingwireless electrical power, at least one switch for coupling at least apart of received wireless electrical power in said at least one firstresonant circuit to at least one load, at least one detector fordetecting a coupling condition of said at least one first resonantcircuit and at least one second resonant circuit, and at least onecontroller for controlling said at least one switch responsive to atleast one of a detected critical- and overcritical coupling condition ofsaid at least one first resonant circuit and said at least one secondresonant circuit.
 17. The wireless power receiver according to claim 16,wherein said at least one first resonant circuit forms a series resonantcircuit and said at least one switch is coupled in series with saidseries resonant circuit and said at least one load, wherein said atleast one switch opens responsive to said at least one of a detectedcritical- and overcritical coupling condition of said at least one firstresonant circuit and said at least one second resonant circuit.
 18. Thewireless power receiver according to claim 16, wherein said at least onefirst resonant circuit forms a parallel resonant circuit and said atleast one switch is coupled in parallel to said parallel resonantcircuit and said at least one load, wherein said at least one switchcloses responsive to said at least one of a detected critical- andovercritical coupling condition of said at least one first resonantcircuit and said at least one second resonant circuit.
 19. The wirelesspower receiver according to claim 18, wherein said at least one switchis placed in parallel to at least one rectifier diode of a rectifier ofsaid at least one first resonant circuit.
 20. The wireless powerreceiver according to claim 16, wherein said at least one detector isresponsive to a period or frequency of at least one of a current andvoltage in said at least one first resonant circuit to detect said atleast one of said critical and overcritical coupling condition.
 21. Thewireless power receiver according to claim 16, further comprising: atleast one of a voltage step-up converter, a voltage step-down converterand a combined voltage step-up/step-down converter implemented betweensaid at least one first resonant circuit and said at least one load. 22.The wireless power receiver according to claim 21, wherein said at leastone switch is part of said at least one of said voltage step-upconverter, voltage step-down converter and combined voltagestep-up/step-down converter.
 23. The wireless power receiver accordingto claim 16, further comprising: at least one voltage- or current loopfor controlling a voltage or current in said at least one load in aclosed loop manner.
 24. The wireless power receiver according to claim16, wherein said at least one first resonant circuit and said at leastone second resonant circuit have substantially equal resonancefrequencies.
 25. A method of receiving wireless power comprising: a)receiving wireless power in at least one first resonant circuit, b)detecting at least one coupling condition of said at least one firstresonant circuit coupled to at least one second resonant circuit, and c)coupling at least a part of said wireless power received to at least oneload via at least one switch, wherein said at least one switch isresponsive to at least one of a detected critical- and overcriticalcoupling condition.
 26. The method according to claim 25, wherein saidcoupling power to said at least one load responsive to at least one of adetected critical- and overcritical coupling condition is at least oneof; c1) disabling coupling of said at least a part of wireless powerreceived to said at least one load via said at least one switch, and c2)limiting coupling of said at least a part of wireless power received tosaid at least one load via said at least one switch.
 27. The methodaccording to claim 25, wherein said at least one first resonant circuitforms a series resonant circuit and said coupling comprises: openingsaid at least one switch responsive to said at least one of a detectedcritical- and overcritical coupling condition, wherein said at least oneswitch is coupled in series with said series resonant circuit and saidat least one load.
 28. The method according to claim 25, wherein said atleast one first resonant circuit forms a parallel resonant circuit andsaid coupling comprises: closing said at least one switch responsive tosaid at least one of a detected critical- and overcritical couplingcondition, wherein said at least one switch is coupled in parallel tosaid parallel resonant circuit and said at least one load.
 29. Themethod according to according to claim 28, wherein said at least oneswitch is placed in parallel to at least one rectifier diode of arectifier of said at least one first resonant circuit.
 30. The methodaccording to claim 25, wherein said detecting is responsive to a periodor frequency of at least one of a current and voltage in said at leastone first resonant circuit to detect said at least one of said criticaland overcritical coupling condition.
 31. The method according to claim25, wherein said coupling via said at least one switch comprises atleast one of a voltage step-up converter, a voltage step-down converterand a combined voltage step-up/step-down converter implemented betweensaid at least one first resonant circuit and said at least one load. 32.The method according to claim 25, further comprising: controlling atleast one voltage- or current in said at least one load in a closed loopmanner.